Electronical amplitude modulator, in particular for modulating signals intended for navigation purposes

ABSTRACT

An electronic amplitude modulator for modulating low frequency signals on the radio or high frequency signals from a transmitter. The associated antenna system can be supplied with the signals which are required according to the conventional structure of the system. Means are provided for bringing the amplitude of the output signal to a number of predetermined values having such duration and being in such succession that the desired envelope of the output signal is obtained with sufficient approximation. Such means comprise a transmission line adapted to be short-circuited electronically by means of shorting devices located at preselected fixed points along the line.

United States Patent [191 Bakken [75] Inventor: Petter Magnar Bakken, Trondheim,

Norway [73] Assignee: Elektronikklaboratoriet ved NTH, Gloshaugen, Trondheim, Norway [22] Filed: July 31, 1972 [21] Appl. No.: 276,800

Related US. Application Data [63] Continuation-impart of Ser. No. 28,788, April 15,

1970, abandoned.

[30] Foreign Application Priority Data ['58] Field of Search 3'43/107, 109; 332/44, 48, 332/42, 31 R, 46; 325/135, 446

[56] References Cited UNITED STATES PATENTS 2,697,220 12/1954 Hancock 343/107 Shortlnq Diode Mauns 4/1970 Perkins 343/109 X l/1971 Hofgen 332/44 FOREIGN PATENTS OR APPLICATIONS 118,707 7/1944 Australia 343/107 1,134,540 11/1968 Great Britain ..332/44 Primary ExaminerAlfred L. Brody Attorney, Agent, or FirmWenderoth, Lind & Ponack 5 7 ABSTRACT An electronic amplitude modulator for modulating low frequency signals on the radio or high frequency signals from a transmitter. The associated antenna system can be supplied with the signals which are required according to the conventional structure of the system. Means are provided for bringing the amplitude of the output signal to a number of predetermined values having such duration and being in such succession that the desired envelope of the output signal is obtained with sufficient approximation. Such means comprise a transmission line adapted to be shortcircuited electronically by means of shorting devices located at preselected fixed points along the line.

6 Claims, 4 Drawing Figures CSB MODULATOR SBO PAYENTEDAU: 13 I974 SHEET 1 BF 3 mmU ATTORNEYS PATENTEDAUGISIBN 3829.796

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INVERTING/ AMF.

INVENTOR PETTER MAGNAR BAKKEN BY I 4 7/JWMKWM M ATTORNEYS PATENTEDAUB 1 3 I974 SHEET 3 BF 3 INVENTOR PETTER MAGNAR BAKKEN AT TOR N E'YS ELECTRONICAL AMPLITUDE MODULATOR, IN PARTICULAR FOR MODULATING SIGNALS INTENDED FOR NAVIGATION PURPOSES This application is a continuation-in-part of application Ser. No. 28,788 filed Apr. 15, 1970, now abancloned.

An important part of localizer equipment for instrument landing systems (ILS) is played by the modulator. This modulator modulates low-frequency signals of 90 and 150 Hz, respectively, on the radio or highfrequency signal from the transmitter, so that the associated antenna system can be supplied with those signals which are required according to the conventional structure of such systems.

The object of the present invnetion is to provide an amplitude modulator for the above purpose. However, it will be realized that the modulator according to the invention can also be used for modulating signals for other navigation purposes.

Among the requirements posed'on modulators for such purposes mentioned shall, in particular, be made of the strict requirements as to phase-shift through the modulator and to the balance between the depth of modulation of the two low-frequency signals. The requirements with respect thereto are far more strict than those being posed on modulators for general communication purposes.

In order to obtain sufficient precision and long term stability, it has heretofore been common to use mechanical modulators for these purposes. Such mechanical modulators are based on the provision of rotors which are rotated with a constant rotational velocity and which modulate the power by introducing capacity changes in electric circuits. An example of such conventional mechanical modulators is described in Norwegian Patent No. 85,874.

There has been a desire to eliminate the conventional modulators based on mechanical movement, which, among others, involve disadvantages with respect to undesired modulator frequencies in particular the frequencies of 30 and 60 Hz and cross-modulation. The rotating mechanical parts are moreover regarded as less than satisfactory with respect to operational reliability and maintenance.

There has therefore been developed electronic modulators which are independent of moving parts. British Pat. Nos. 1,134,540 and 1,174,070 are related to suchv electronic modulators. These modulators comprise means which cause the amplitude of the output signal to assume a number of predetermined values with such duration and in such succession that the desired envelope of the output signal is obtained with sufficient approximation.

Reference is also made to the book Electronic Navigation Engineering by Peter C. Sandretto, 195 8, discussion on page 641, an electronic ILS modulator based on conventional AM-anode modulation. Reference is further made to an article by J. Stammelbach in International Elektronische Rundschau, 1968 Nr. 7, and an article by G. Hofgen in Frequenz Nr. 22, 1968, describing a modulator function based on the continuously variable RF-resistance in a diode. Modulators according to these principles are not sufficiently accurate for [L8 applications.

Thus, by making the envelope of the output signal assume the shape of a step-curve, advantage is taken of the fact that the requirements as to the distortion factor in said envelope are comparatively moderate according to the international rules applying to instrument landing systems. According to these rules, up to 10 percent distortion is permitted in the envelope. When a sinusoid wave is approximated by a step-curve with constant step-height, the distortion factor k is determined by the expression:

k z Vl76' l/N' 100% where N is the number of steps per half cycle, which means that the total number of steps is 2N l-l. If 15 steps are used, the distortion factor will be:

k z V176-l/7-l00=5.8%

The function of modulators according to this principle can be determined by passive components. This has the advantage that the long-term stability is very good and that the modulation will be independent of the power level in the carrier wave.

It is an object of this invention to provide an improved and simplified electronic modulator of the above type. The present invention is directed to a modulator with a more effective utilization of the shorting diode and which more simply enables the adjusting of the percentage of modulation.

The means in the modulator for producing the steplike envelope of the output signal according to the ,invention comprises at least one transmission line adapted to be short-circuited electronically by means of shorting devices located in preselected fixed points along the line.

By having the transmission lines in this way, high stability of both the desired high-frequency or carrier wave phase locking and the degree of modulation are obtained. This is of great importance in localizer equipment for instrument landing systems.

The length of the transmission line or lines is varied I by means of said electronic short-circuiting, for in stance, in the form of diodes which are controlled by direct current pulses from switching circuits provided for that purpose.

By applying a carrier wave signal to both ends of the transmission line, which signal is reflected at the diode shorted at that instance, the phase shifts of the reflected signals are accurately related to one another and can be made equal and opposite.

A more detailed explanation of the invention and possible embodiments thereof, as well as further particular features, will be given in the following description referring to the drawings, in which:

FIG. 1 shows a simplified diagram of a modulator according to a first embodiment of the invention;

FIG. 2 shows a corresponding diagram of a second embodiment according to the invention;

FIG. 3 shows a block diagram of control and switching circuits for the transmission lines in the modulator according to the invention; and

FIG. 4 shows a portion of a simplified circuit diagram of the switching circuit.

The diagram of FIG. I shows a structure corresponding to the conventional arrangement for modulating a high-frequency or carrier wave signal RF being supplied from the associated transmitter. It contains an input hybrid 1 with terminating resistor 3, two modulators 4 and for modulating 90 and 150 Hz, respectively, and an output hybrid 2 with outputs CSB and SBO for the combination of the carrier wave and the side bands, and the side bands alone, respectively. These two outputs lead the respective signals to a distribution network which distributes the signals to the separate antennas in the associated antenna system. The subsequent handling of the signals from the two outputs CSB and SEC in the distribution network and in the antenna system may take place in the conventional way, and therefore will not be explained further here.

Whereas the two modulators in the structures applied hitherto for use in instrument landing systems have been based on mechanically movable parts, the modulators 4 and 5 applied in connection with the present invention are built with purely electronic components. In FIG. 1, only the modulator 4 is shown with its separate components, whereas the modulator 5 is indicated as a block, since the same is built in exactly the same way as the modulator 4.

In the input hybrid 1 the carrier wave from the transmitter is split into two like portions which are fed each to one of the two modulators 4 and 5. The modulated signals V and V respectively, being provided by the modulators are applied to the output hybrid 2 which, in the known way, combines these signals to produce signals CSB and SEC.

The function of the modulator 4 in FIG. 1 is based on a splitting of the input signal to the hybrid 6 into two signal paths. Each path has a phase-shifter which is adapted to provide a phase shift equal in magnitude to that of the other path, but in the opposite sense. The oppositely phase shifted signals are thenapplied to a hybrid 7 which combines the signals to a useful output signal V and an unuseful output signal V the latter being absorbed in the terminating resistor 11. Said opposite phase shifts are indicated in FIG. 1 by the respective angles +4: and -11). The phase difference between the two signals which are applied to the input of the hybrid 7, accordingly, will be 2d), and it will be realized that said signals V, and V will have the following form:

In the above equations V is the voltage applied to the hybrid 7. It will be seen that the phase of the output voltages V and V is independent of (11, whereas the amplitude is dependent upon (1:. This, then, makes it possible to obtain an amplitude modulation without affecting the phase.

The two phase-shifters in the modulator 4 each com With the arrangement just described, each phase shifter will produce a step-wise variation of the phase angle d), and thereby of the amplitude of the signal V The short-circuiting of the two transmission lines 12 and 13 is controlled in coordination, and the electrical length of the lines are seen, for instance from the respective ports 14 and 15 of the hybrid 8, is so adjusted that there is a phase difference of M4 between the two lines. This means that all of the power being fed to the hybrid 8 will be supplied therefrom to the hybrid 7 when the waves are reflected from the respective shortcircuiting locations in the transmission lines 12 and 13. If the short-circuit is moved an electrical length 6, the propagation length will change by two 0 in each of the lines, as seen from the respective ports 14 and 15, so that ()5 20. The phase-shifter being built around the hybrid 9 functions in a completely analogous way. Since the phase shifting in the two signal paths shall be opposite and of equal magnitude H111 and respectively), the same short-circuits in the transmission lines 12 and 13 can be used for both phase shifters, the waves from each of the hybrids 8 and 9 approaching the short-circuits from different sides thereof. The embodiment shown in FIG. 1 thus enables an effective utilization of the transmission lines and the associated short-circuiting devices, thus, among other things, reducing the number of short-circuiting devices which are needed. An at least equally important advantage, however, is the fact that in this way the variation of the phase-shift (I) will necessarily be equal and opposite for the two signal paths or branches.

It should be noted that the modulators 4 and 5 in FIG. 1 are matched during the complete modulation period, i.e., during the complete cycle of the lowfrequency signals, so that the cross-modulation will be zero.

The short-circuits are, in a practical embodiment, effected by means of silicon rectifier diodes which in the open condition are backwardly biased with 270V, and in the short-circuiting condition are supplied with a conducting current of about 0.1A.

In FIG. 1 the input to hybrid 6 is the radio frequency input means of the modulator 4, whereas the right-hand port of hybrid 7 serves as the output means fromwhich the modulated output signal is delivered.

In FIG. 2, which shows a second embodiment of the modulator according to the invention, the conventional components, as shown also in FIG. 1, are found, i.e. the input hybrid l with terminating resistor 3 and the output hybrid 2 with outputs CSB and SEC. The arrangement of FIG. 2 utilizes the recently developed nonreciprocal components, isolators or circulators for the frequencies concerned here. With such components it is apparent from FIG. 2 that it is possible to simplify the structure of the arrangement to a large degree compared with that shown in FIG. 1. Accordingly, the modulator of FIG. 2 is regarded as preferable to the one in FIG. 1.

The circulators 24 and 25 with their associated terminating resistors 26 and 27 operate in the known manner to allow the unhampered transmission of power from the hybrid l to the respective hybrids 21 and 22 in the two modulators. However, power coming from these modulator hybrids (for instance as represented by the signal V, from the hybrid 21) is led to the terminating resistors of the circulators and will not revert to the input hybrid l.

The modulator which is built around the hybrid 21 and which can, for instance, be adapted to effect the modulation of the low-frequency signal 90 Hz, employs only one single transmission line 23 with a number of short-circuiting points which affect the waves arriving thereat from both sides, i.e., from the respective ports P and P of the hybrid 21. Thus, the input to circulator 24 constitutes the radio frequency input means of the 90 Hz modulator and the output means of the modulator is port P of hybrid 21. A more detailed explanation of the operation is as follows.

A wave arriving at port l will be distributed on ports P and P After having passed through the lines with electrical length qb and (1) as indicated in FIG. 2, the waves are reflected from the short-circuiting point or plane and impinge once again on the ports P and P If the voltage amplitude of the two equal and reflected waves is designated V the voltage across port I" will be as follows:

bi 1 2 be d), (b 1r/2 0 (1) d) IT/2 6 V2: V0 1/ J( 0 w/z +0 i 1:12 o V =V Vie -cos0 In a corresponding way, it will be seen that the reflected voltage to the input at port P will be:

V V V2 e sin 6 The power due to the last-mentioned signal voltage is absorbed in the load or terminating resistor 26 of the circulator 24. This is necessary so as to obtain impedance matching at the input which results in low crossmodulation.

From equation (6) it appears that constant phaseshift through the modulator is obtained as a consequence of the fact that (1) is a constant. This is accomplished by employing only one single transmission line in connection with the hybrid so that corresponds to the total geometric length of this line.

With non-perfect short-circuiting there will be a certain leakage of power past the diodes in the transmission line, but this power will be absorbed in the terminating resistor of the circulator and therefore will not detract from the output signal.

The transmission lines applied in FIGS. 1 and 2 can be replaced by artificial lines in the case of transmitters working at lower frequencies, for instance in VHF radio beacons. By effecting changes in certain line lengths, the modulator according to the invention can also be used as a balanced modulator which can likewise be suitable for navigation purposes.

The hybrids shown in FIGS. 1 and 2 can be replaced by quadrature hybrids in which the outgoing signals have a phase difference of Arr/2.

In addition to the alternatives illustrated in FIGS. 1 and 2, it is possible within the scope of this invention to make a third embodiment by employing only one circulator located in the transmitter output. Also, this embodiment will give low cross-modulation, provided that the circulator has a low standing wave ratio at the output port.

The short-circuiting devices, for instance the diodes in the above-described transmission lines, will, as mentioned, need control pulses for the shorting thereof in accordance with the instantaneous value of the modulating low-frequency signal. The necessary control circuits and switching circuits for this purpose are shown in FIG. 3 and a portion of the switching circuit is shown in more detail in FIG. 4. In FIG. 3, there is shown a 900 Hz oscillator 30, the output of which is connected to the input of two frequency dividing circuits 31a and 31b which effect a frequency division by six and 10, respectively, whereby the desired low-frequencies of 150 and 90 Hz are generated. In low pass filters 32a and 32b, the low-frequency signals are filtered in order to obtain the desired sinusoid shape thereof. By generating in this way the desired low-frequency signals from a common fundamental frequency of 900 Hz, it is possible to obtain the necessary mutual phase locking between the low frequency signals. As a further aid in accomplishing this phase locking, the output signal from the dividing circuit 31b can be further divided by 3 and via a monostable vibrator be applied to a reset input of the circuit 31a, so that the same at the beginning of each working period is forcibly set identically with the circuit 31b. This modification is not shown in detail in FIG. 3, but is only indicated with the arrow 38.

The filtered low-frequency signal of 150 Hz is fed from the low-pass filter 32a directly to a switching circuit 36a and is also applied to a rectifier 33a which is followed by a potentiometer 39a and then by a low-pass filter 34a which delivers a smoothed DC voltage to the switching circuit 36a. This smoothed DC. voltage is also applied to the switching circuit 36a through an inverting amplifier 35a with amplification equal to l. The DC. voltage from the low-pass filter 34a represents a reference voltage which is made equal to the peak value of the low frequency voltage by means of a precision half wave rectifier with a built in amplification equal to 2 V2 In a practical embodiment of the rectifier, an operational amplifier can be used so as to obtain good temperature stability.

An important feature of the control circuits shown is the potentiometer 39a which makes it possible to regulate or adjust the said amplification, thus making possible corresponding adjustment of the degree of modulation.

The circuits which handle the low-frequency signal of i 90 Hz from the low pass filter 32b, i.e. the circuits 33b,

39b, 34b and 35b, correspond exactly to the above described circuits for handling the low frequency signal of Hz, and therefore should not need further explanation. Moreover, the corresponding two switching circuits 36a and 36b are quite similar.

Each of these switching circuits has 15 output pairs, each output of the pair being identical to the other and each pair representing the output of one switching unit. The circuits are built up of 15 such units.

The exact time of switching on the various diodes for shorting thereof in the desired succession timed with the variations of the low frequency signal concerned is determined by a comparison between the sinusoid voltage of the low-frequency signal and a number of reference levels or steps. As already mentioned, this method, among other things, has the advantage that the degree of modulation can be varied by varying the ratio between the amplitude of the sinusoid voltage and the reference D.C. voltage.

The reference voltage or voltages (positive and negative) are sub-divided in each switching circuit into portions or steps which determine at which instantaneous value of the low frequency signal the switching over from one shorting diode to the next takes place. An example of a practical circuit for performing this is shown in FIG. 4. This figure shows a portion or section comprising three units or channels designated with numerals 80, 90 and 100. These are adapted to control the respective corresponding diodes numbered 80, 90 and 100 on a transmission line. The points designated S in FIG. 4 are connected together and to a low frequency signal, i.e. either 90 Hz or 150 Hz is applied thereto.

Referring to FIG. 4, the operation of a switching circuit is as follows. Assume that the signal at the points S has had its largest negative value. When the instantaneous value of the signal at S reaches the reference voltage at the base of transistor Q the current from the current generator transistor Q passes through transistor Q and a signal is produced at transistors Q and Q and at the diodes. At the same time, the transistors Q and Q are blocked and this in turn results in the blocking of transistor Q so that the pulse at the output of channel 80 disappears.

When the low frequency voltage becomes equal to V -3/ 14, the transistors Q and Q are blocked and the same will, accordingly, be the case for transistor G whereby the signal in channel 9 is eliminated. Transistor O becomes conducting and an output signal is established in channel 100. The current generator transistor (not shown) for channel 150, i.e. @151 applies current continuously. The latter transistor is situated in a corresponding location as the respective transistors 0 0 0 etc. In channel 150 there is thus no transistor corresponding to the respective transistors Q Q 0 and so on.

Referring to FIG. 3, it will be obvious to a person skilled in the art that the reference voltage can be gen erated in other ways than by rectification of the respective low frequency signals. In such case, a stabilized voltage source for the reference .voltage would be necessary. It is, however, very advantageous to let the reference voltage be dependent upon the low-frequency signal because this eliminates errors and inexactitudes due to possible mutual independent variations of the amplitude of the low frequency signal and the reference voltage. It will further be obvious that the control circuits can be built in various other ways which also employ conventional components and techniques.

What is claimed is:

1. Electronic amplitude modulator producing a modulator output signal intended for navigation purposes, in particular for use in instrument landing systems (ILS), comprising:

radio frequency input means;

means connected to said input means for causing the amplitude of the modulator output signal to assume a number of predetermined values of such duration and in such succession that a desired envelope of modulated output signal is obtained with sufficient approximation, said means comprised of at least one transmission line, electronically controlled shorting means located at preselected fixed points along said at least one transmission line and being adapted to short-circuit said transmission line electronically, for varying the amplitude of said output signal, and

means applying radio frequency signals from said radio frequency input means to both ends of said at least one transmission line, whereby said shorting means act on waves impinging thereupon from both directions; and

hybrid circuit means operatively connected in the modulator having a plurality of ports, one of said ports serving as output means from which the modulated output signal is delivered.

2. The amplitude modulator according to claim 1 wherein two transmission lines are used in said means for causing the amplitude of the modulator output signal to assume a number of predetermined values and wherein said means applying radio frequency signals at both ends of said transmission lines is comprised of first and second hybrid circuit means, each having two oppositely disposed ports, one of said two transmission lines being connected between one of said oppositely disposed ports of each of said first and second hybrid circuit means, and the second of said transmission lines being connected between the other of each of said oppositely disposed ports of said first and second hybrid circuit means.

3. An amplitude modulator according to claim 1 wherein said shorting means are semiconductor diodes having a low RF impedance value and a high RF impedance value depending on a D.C. bias applied to said diodes, D.C. bias being supplied thereto in the form of direct current pulses.

4. An amplitude modulator as recited in claim 3 wherein means are provided for forming a reference D.C. voltage from a low frequency input signal source and switching circuit means connected to said means for forming a reference D.C. voltage, the low frequency input signal source and said semiconductor diodes for delivering direct current pulses to said semiconductor diodes for modulation of said radio frequency signal,

and wherein means for adjusting said reference D.C.

voltage are operatively connected between said means for forming a reference D.C. voltage and the low frequency input signal source for setting the predetermined values of the amplitude of the modulated output signal.

5. Amplitude modulator according to claim 1 wherein said means applying radio frequency signals at both ends of said transmission line comprises:

a hybrid circuit means having a pair of oppositely disposed ports,

said transmission line being connected between said ports, and

a circulator between said radio frequency input means and said hybrid circuit for feeding radio frequency power to, and for absorbing unwanted power reflected from, said hybrid circuit.

6. Amplitude modulator according to claim I wherein said transmission lines are formed by artificial lines. 

1. Electronic amplitude modulator producing a modulator output signal intended for navigation purposes, in particular for use in instrument landing systems (ILS), comprising: radio frequency input means; means connected to said input means for causing the amplitude of the modulator output signal to assume a number of predetermined values of such duration and in such succession that a desired envelope of modulated output signal is obtained with sufficient approximation, said means comprised of at least one transmission line, electronically controlled shorting means located at preselected fixed points along said at least one transmission line and being adapted to short-circuit said transmission line electronically, for varying the amplitude of said output signal, and means applying radio frequency signals from said radio frequency input means to both ends of said at least one transmission line, whereby said shorting means act on waves impinging thereupon from both directions; and hybrid circuit means operatively connected in the modulator having a plurality of ports, one of said ports serving as output means from which the modulated output signal is delivered.
 2. The amplitude modulator according to claim 1 wherein two transmission lines are used in said means for causing the amplitude of the modulator output signal to assume a number of predetermined values and wherein said means applying radio frequency signals at both ends of said transmission lines is comprised of first and second hybrid circuit means, each having two oppositely disposed ports, one of said two transmission lines being connected between one of said oppositely disposed ports of each of said first and second hybrid circuit means, and the second of said transmission lines being connected between the other of each of said oppositely disposed ports of said first and second hybrid circuit means.
 3. An amplitude modulator according to claim 1 wherein said shorting means are semiconductor diodes having a low RF impedance value and a high RF impedance value depending on a D.C. bias applied to said diodes, D.C. bias being supplied thereto in the form of direct current pulses.
 4. An amplitude modulator as recited in claim 3 wherein means are provided for forming a reference D.C. voltage from a low frequency input signal source and switching circuit means connected to said means for forming a reference D.C. voltage, the low frequency input signal source and said semiconductor diodes for delivering direct current pulses to said semiconductor diodes for modulation of said radio frequency signal, and wherein means for adjusting said reference D.C. voltage are operatively connected between said means for forming a reference D.C. voltage and the low frequency input signal source for setting the predetermined values of the amplitude of the modulated output signal.
 5. Amplitude modulator according to claim 1 wherein said means Applying radio frequency signals at both ends of said transmission line comprises: a hybrid circuit means having a pair of oppositely disposed ports, said transmission line being connected between said ports, and a circulator between said radio frequency input means and said hybrid circuit for feeding radio frequency power to, and for absorbing unwanted power reflected from, said hybrid circuit.
 6. Amplitude modulator according to claim 1 wherein said transmission lines are formed by artificial lines. 